Quadrature modulator and calibration method

ABSTRACT

A quadrature modulator and a method of calibrating same by applying a first test tone signal to an in-phase modulation branch input of the modulator and a ninety degree phase-shifted version of the first test tone signal to a quadrature modulation branch input of the modulator. The carrier leakage level in an output signal of the modulator is measured and in response base band dc offset voltages are adjusted to minimize the carrier leakage. A second test tone signal is applied to the in-phase modulation branch input and a ninety degree phase-shifted version of the second test tone signal to the quadrature modulation branch input. The level of an undesired upper sideband frequency component in the output signal is measured and in response base band gains the in-phase and quadrature modulation branches and a local oscillator phase error are adjusted to minimize the undesired sideband.

CROSS REFERENCE TO RELATED APPLICATION

This application claims the benefit of U.S. Provisional Application Ser.No. 60/465,127, filed Apr. 24, 2003 which is incorporated herein byreference.

The invention relates to a quadrature modulator, a radio communicationdevice comprising a quadrature modulator, and a method of calibratingthe quadrature modulator or radio communication device.

The invention relates to a quadrature modulator, a radio communicationdevice comprising a quadrature modulator, and a method of calibratingthe quadrature modulator or radio communication device.

In a quadrature modulator, carrier leakage reduction and side bandrejection have typically been carried out using one of two conventionalmethodologies. One of these methodologies depends on circuit matching,dynamic swapping, and the use of polyphase filters, all of which arecarried out in the design phase. The second type of methodology dealswith an imperfect chip but relies on user calibration methods while thechip is in use.

One example of the second methodology is disclosed in U.S. Pat. No.6,169,463 to Mohindra et al. which discloses a quadrature modulator withset and forget carrier leakage compensation. Upon the power-up of thequadrature modulator, carrier leakage is measured in the in-phase andquadrature branches through the use of a synchronous detector. A statemachine starts signal generators which inject compensation signals intothe in-phase and quadrature branches, respectively, so that DC offsetsin these branches are reduced thereby reducing the carrier leakage.Mohindra does not, however, disclose the rejection of undesired sidebands. Mohindra proposes a simple detection scheme which can bedifficult to implement since it does not distinguish between carrierleakage and side-tones or other errors. Also the large dynamic range ofthe signals that have to be suppressed require cumbersome and slow“gain-set” and “error-detect” sequences that have to be implemented insoftware.

It is desirable to not only suppress the carrier leakage but also toreduce undesired sidebands using minimal, and low cost, circuitry. It isalso desirable to minimize the search for appropriate circuit parametersto reduce these undesirable circuit parameters.

One aspect of the invention relates to a method of calibrating aquadrature modulator. The method includes: applying a first test tonesignal to an in-phase modulation branch input of the modulator and aninety degree phase-shifted version of the first test tone signal to aquadrature modulation branch input of the modulator, measuring the levelof a local oscillator (LO) feedthrough in an output signal of themodulator and in response adjusting base band dc offset voltages tominimize the LO feedthrough; applying a second test tone signal to thein-phase modulation branch input and a ninety degree phase-shiftedversion of the second test tone signal to the quadrature modulationbranch input; and measuring the level of an undesired upper sidebandfrequency component in the output signal and in response adjusting baseband gains the in-phase and quadrature modulation branches and a LOphase error to minimize the undesired side band.

In the preferred embodiment, the level of the local oscillator (LO)feedthrough or the undesired sideband in the output signal is measuredby: shifting the frequency spectrum of the output signal such that alower sideband frequency component (LSB) is down-converted to zero IF;filtering the spectrum-shifted signal to pass through either the LOfeedthrough or the undesired sideband, such as an upper sideband; andmeasuring the amplitude of the filtered, spectrum-shifted signal.

Another aspect of the invention relates to a quadrature modulator whichincludes an in-phase modulation branch and a quadrature modulationbranch. The in-phase modulation branch receives an analog in-phase baseband signal as an input, and includes a first dc offset adjustmentcircuit, a first base band gain adjustment circuit, and a first mixer.The quadrature modulation branch receives an analog quadrature basedband signal as an input, and includes a second dc offset adjustmentcircuit, a second base band gain adjustment circuit, and a second mixer.A local oscillator means provides a local oscillator signal to the firstmixer and a phase shifted version of the local oscillator signal to thesecond mixer. A summer sums the outputs of the first and second mixers.An envelope detector detects an output signal of the modulator andprovides a signal representative of the amplitude of the output signal.A band pass filter filters the amplitude signal. A signal strengthindicator circuit measures the strength of the filtered amplitudesignal, and provides a compensation signal for adjusting the phase shiftof the local oscillator and the dc offsets and base band gains of thein-phase and quadrature base band signals.

In the preferred embodiment the envelope detector is a synchronousdetector and the signal strength indicator is a log indicator. Means areprovided for applying a first test tone signal to the in-phasemodulation branch input and a ninety degree phase-shifted version of thefirst test tone signal to the quadrature modulation branch input. Thecompensation signal is employed to minimize carrier leakage in theoutput signal by adjusting the base band dc offsets in the in-phase andquadrature branches. A second test tone signal is then applied to thein-phase modulation branch input and a ninety degree phase-shiftedversion of the second test tone signal is applied to the quadraturemodulation branch input. The second test tone has a frequency that issubstantially one half of the frequency of the first test tone. Thecompensation signal to is employed minimize an undesired upper sidebandfrequency component in the output signal by adjusting the base bandgains the in-phase and quadrature modulation branches and the phaseshift of the local oscillator signal.

The preferred synchronous envelope detector comprises a Gilbert cellhaving at least one differential transistor pair in an upper branch andat least one transistor in a lower branch, the upper and lower branchesbeing interconnected, with each of the upper and lower branches havinginput terminals. A resistor divider network is connected between theinput terminals of the upper branch and the input terminals of the lowerbranch. The resistive values of the network are selected such that aselected input signal having a signal level sufficient to saturate thetransistors of the upper branch is attenuated so as to not saturate thetransistors of the lower branch. A low pass filter is connected to theupper branch of transistors, and an output signal of the detector isprovided at the low pass filter.

FIG. 1 is a system block diagram of a prior art quadrature modulator.

FIG. 2 illustrates the existence of various non-idealities in thequadrature modulator of FIG. 1.

FIG. 3 illustrates the waveform of a two-tone signal in the time domain.

FIG. 4 illustrates the output spectrum of the prior art quadraturemodulator in the frequency domain when the base band input is a testtone.

FIG. 5 is a system block diagram of a quadrature modulator according tothe preferred embodiment which includes circuitry for calibrating themodulator.

FIG. 6 illustrates the output spectrum of the preferred quadraturemodulator in the frequency domain when a first test tone is appliedthereto for the purposes of a calibration phase to eliminate carrierleakage;

FIG. 7 illustrates the output spectrum of the preferred quadraturemodulator in the frequency domain when a second test tone, being onehalf the frequency of the first test tone, is applied to the modulatorfor the purposes of a second calibration phase to eliminate an undesiredside band;

FIG. 8 is a circuit diagram for an envelope detector employed in thepreferred embodiment.

FIG. 9 is a circuit diagram for a signal strength indicator and bandpass filter employed on the preferred embodiment.

FIG. 1 shows a quadrature transmitter 8 which includes an in-phasemodulation branch 10 and a quadrature modulation branch 12. The in-phasebranch 10 includes, in series arrangement, a base band dc offsetadjustment block 16, a low pass filter 18, a base band gain adjustmentamplifier 20, and mixer 22. The mixer 22 mixes an in-phase signal I(t)(an analog signal carrying digital information) with a sinusoidalcarrier signal A_(c) cos(ω_(t)) generated by a local oscillator (LO) 24.The quadrature phase branch 12 includes, in series arrangement, a dcbase band offset adjustment block 17, a low pass filter 19, a gainadjustment amplifier 21, and a mixer 23. The mixer 23 mixes a quadraturesignal Q(t) with a carrier signal A_(s) sin(ωt+φ_(e)) that is generatedby the local oscillator 24 and phase shifted 90°, in the ideal case, bya phase shift circuit such as a phase locked loop (PLL) 26. φ_(e)represents the phase shift error.

The outputs of the mixers 22 and 23 are summed by a summer 28, theoutput of which is fed to a digitally programmable attenuator 30. In thepreferred embodiment the transmitter 8 operates on differential signals,and so the output of the attenuator 30 is fed to a transformer 32 whichconverts the differential signals to a single ended signal to beradiated through an antenna 34.

A digital signal processor (DSP) 36 controls various circuits such asthe DC offset adjustment blocks 16, 17, the gain amplifiers 20, 21, thePLL 26 and the attenuator 30. The DSP 36 also executes a calibrationalgorithm which is described in greater detail below.

A number of imperfections or non-idealities are present in thequadrature transmitter 8 as shown, and in other quadrature transmitterssimilar thereto. These non-idealities arise, inter alia, from thefollowing sources, as indicated in FIG. 2:

-   -   A_(I), A_(Q) Baseband gain in the in-phase and quadrature        branches, respectively, which may not be equal    -   V_(OI), V_(OQ) Built-in equivalent dc offset voltage in the        in-phase and quadrature branches, respectively, which may not be        zero    -   A_(C), A_(S) Local oscillator amplitudes in the in-phase and        quadrature branches, respectively, which may not be equal    -   V_(OC), V_(OS) Local oscillator equivalent offset voltages in        the in-phase and quadrature branches, respectively, which may        not be zero    -   φ_(e) Local oscillator quadrature phase error

Consequently, the transmit signal, s(t), can be written in the followinggeneral form:s(t)=(A _(I) I(t)+V _(OI))(A _(C) cos(ω_(LO) t)+V _(OC))+(A _(Q) Q(t)+V_(OQ))(A _(S) sin(ω_(LO) t+φ _(e))+V _(OS))  [1]

where

φ_(LO) is the local oscillator frequency, and

I(t), Q(t) are the in-phase and quadrature signals, as discussed above.

It will be seen from equation [1] that if the non-idealities areeliminated, s(t) reduces to an ideal quadrature signal:I(t)cos(ω_(LO)t)+Q(t)sin(ω_(LO)t)  [2]

If the input baseband signals are defined as I(t)=V_(I) cos(ω_(B)t) andQ(t)=V_(Q) sin(ω_(B)t), expanding equation [1] yields three major termsas follows:V_(OI)*A_(C) cos(ω_(LO)t)+V_(OQ)*A_(s) sin(ω_(LO)t+φ_(e))  [3]A_(I)V_(I)A_(C) cos(ω_(LO)t−ω_(B))−A_(Q)V_(Q)A_(S)cos(φ_(e))cos(ω_(LO)t+ω_(B)t)+A_(Q)V_(Q)A_(S)sin(φ_(e))sin(ω_(LO)t+ω_(B)t)  [4]A_(I)V_(I)A_(C) cos(ω_(LO)t−ω_(B)t)+A_(Q)V_(Q)A_(S)cos(φ_(e))cos(φ_(LO)t−φ_(B)t)−A_(Q)V_(Q)A_(S) sin(ω_(LO)t−ω_(B)t)  [5]

Equation [3] is referred to as the local oscillator (LO) feed-through orcarrier leakage term because this component of the output spectrum iscentered at ω_(LO) It will be seen that if the dc offset adjustment canbe properly adjusted at blocks 16 and 17, the LO feed-through can beeliminated or reduced.

Equation [4] is referred to as the “upper side band” (USB) term sincethis component of the output spectrum is centered at a frequency,ω_(LO)+ω_(B), which is higher than the local oscillator frequency.Similarly, equation [5] is referred to as the “lower side band” (LSB)term since this component of the output spectrum is centered atω_(LO)−ω_(B), which is lower than the local oscillator frequency.

Assuming positive parameters, the LSB will be the stronger frequencycomponent and hence the desired signal, and the USB will be the weaker,and hence undesired signal. From equation [4], it will be seen that abase band gain mismatch, A_(I)≠A_(Q), a local oscillator level mismatch,A_(C)≠A_(S), and the local oscillator phase error φ_(e) are responsiblefor the residual USB term. The proper control of these system parameterscan minimize the undesired sideband. However, efforts to minimize theundesired sideband to less than 50 dBc in the analog domain requiresbase band and high frequency gain matching on the order of 0.05 dB andphase matching of 0.4 degrees. Conventional analog circuits do not allowsuch matching levels to be achieved.

Rather than trying to improve matching and phase control by eitherimproved layouts or dynamic techniques, the present invention measuresthe output signal s(t) and adjusts the following system parametersA_(I), A_(Q); V_(OI), V_(OQ); and φ_(e) until the LO feed-through andUSB components are minimized. This is accomplished by taking advantageof the fact that the LSB component will typically have a greatermagnitude than the USB component or the LO feed-through. FIG. 3exemplifies a two tone signal 40 in the time domain which is a sum oftwo sinusoidal components, A_(St) cos(ω_(st)t)+A_(we) cos(ω_(we)t),where the first component has a much higher amplitude than the secondcomponent. It will be seen from FIG. 3 that the envelope 42 (shown instippled lines) of the two tone signal 40 is a sinusoid having theamplitude of the weak signal and a frequency equal to the differencebetween the two tones. More particularly, the envelope of the two tonesignal 40 can be written as A_(St)+A_(we) cos(ω_(st)t−ω_(we)t). Thus,if, for example, the weaker signal is the USB component and the strongersignal is the LSB component, measuring the envelope of such a two tonesignal will enable the opportunity to adjust system parameters to reducethe level of the weaker or undesired signal.

In practice, however, the output s(t) of the transmitter, is not a twocomponent signal but will have many frequency components. FIG. 4illustrates a typical frequency spectrum for s(t). In this plot, thecentral axis is centered on the local oscillator frequency, e.g., at 1GHz. Frequency component 44 represents the LO feed-through, component 46represents the USB and component 48 represents the LSB. In addition,strong and weak 3^(rd) harmonics exist, as represented by components 50and 52, respectively. Accordingly, as the output signal s(t) hasmultiple frequency components in addition to the desired component, thepreferred embodiment filters at least some of the frequency componentsso that a two tone signal is substantially present. Note that the higherharmonics, being relatively small in amplitude, will not materiallyaffect the shape of the envelope. In addition since the higher harmonicsdecrease substantially faster than the main signal the user will be ableto reduce them by proper choice of the main signal levels (e.g. forevery 1 dB reduction in the level of the LSB as in FIG. 4, the 3^(rd)harmonics go down 3 dB).

FIG. 5 is a system block diagram of a quadrature modulator 100 accordingto the preferred embodiment of the invention which employs a highdynamic range envelope detector 102, a band pass filter 104, and asignal strength indicator circuit 106. The detector 102 is positioned todetect s(t) at point A, at the output the attenuator 30. The detector102 provides a signal at point B which represents the envelope of s(t).This envelope is then filtered by the band pass filter 104 so as toallow through (at point C) the frequency component that should beminimized. The logarithmic signal strength indicator circuit 106measures the strength of the passed-through frequency component andprovides a signal 108 which is then used by the DSP 36 to adjust thesystem parameters of the modulator.

The modulator is preferably calibrated In a two phase process. In thefirst phase, the LO feed-through is minimized and in the second phasethe USB component is minimized, although in alternative embodimentsthese phases may be executed reversed in order.

In the preferred first phase, a first test tone, e.g., 4 MHz, is appliedto the I and Q base band inputs of the transmitter (the tone applied atthe Q input being 90 degrees out of phase with the tone applied at the Iinput). The tones are generated by the DSP 36 or alternatively by anyother known tone generator. The output spectrum at point A is as shownin FIG. 4, having the desired LSB component 46, and smaller LOfeed-through and USB components 44, 48 plus 2^(nd) and 3^(rd) harmoniccomponents. The envelope detector 102 essentially shifts the frequencyspectrum of the detected s(t) such that at point A, as shown in FIG. 6,the LSB component 46 is shifted to zero frequency, the LO feed-throughcomponent 44 is shifted to the frequency of the test tone, 4 MHz, andthe USB component 48 is shifted to a frequency that is double that ofthe test tone. The band pass filter 104 is preferably configured with asharp passband centered at 4 MHz which passes through substantially onlythe LO feed-through (as schematically illustrated in FIG. 6) 44, andthus controls the presence of various other undesired components. Thelogarithmic signal strength indicator 106 measures the level of the LOfeed-through 44 and generates a signal which is used by the DSP 36 tominimize the LO feed-through 44. The LO feed-through 44, even though itmay arise as a result of leakage through the substrate of the chip canbe nulled out by adjusting the DC offset voltages V_(OI) and V_(OQ) soas to cause the LO feed-through 44 to cancel out at point A. Note thatV_(OI) and V_(OQ) are independent of each other and need to beseparately nulled, thus requiring a two dimensional search to find theoptimum values for the DC offsets.

In the second phase of the calibration process, once the LO feed-through44 has been minimized, a second test tone at half the frequency of thefirst test tone, e.g., 2 MHz, is applied to the I and Q base band inputsof the modulator. After passing through the envelope detector 102 thefrequency spectrum at point C resembles that shown in FIG. 7, whereinthe USB component 48 is separated from the LSB component 46 by twice thevalue of the second test tone signal. This allows the very same bandpass filter 106 to be used to propagate substantially only the USBcomponent 48 (as schematically illustrated in FIG. 7) to the logarithmicsignal strength indicator 106. The signal strength indicator 106generates the signal 108 which is used by the DSP 36 to adjust the baseband gains A_(I) or A_(S) at gain adjustment blocks 20, 21 and the localoscillator phase error φ_(e) at PLL 26 in order to minimize the USBcomponent. This also requires a two dimensional search.

The calibration process is of the “set and forget” type. It may beapplied upon power up of the modulator, or at certain discrete instancessuch as inactive time slots. There is no need to continuously adjust thesystem parameters.

Note that the band pass filter 104 will reduce the effect of the strong3^(rd) harmonic distortion component 50. The weak 3^(rd) harmoniccomponent 52, however, falls directly on the sideband signal 48 afterthe envelope detector. The level of this distortion product is typicallymore than 60 dBc and is not likely to cause any problems. Its level canbe reduced by 3 dB for every dB of reduction in the main test tonesignals I(t) and Q(t) during the calibration phase.

A bandpass filter can be arranged prior to the envelope detector butthat requires costly high Q filters that are impractical. Those skilledin the art will the band pass filter can be a programmable band passfilter disposed after the detector to filter out undesired sidebands andharmonic products.

In the preferred embodiment the modulator 100 provides a large dynamicrange capacity, for a number of reasons. First, the RF output level mayneed to be programmable, which is why the quadrature modulator 100includes attenuators 30. The detector circuitry preferably detects thesignal after the attenuation to minimize signal non-ideality at thepoint of delivery. Consequently, the level of the LSB component 46 couldchange dramatically.

Second, it is desired to reduce considerably the level of the detectedfrequency component, e.g. the LO-feed-through or USB component 48, e.g.from 15 dBc as a starting point to 50 dBc. Adding the variation of themain signal due to the RF level adjustments in the output attenuator,e.g. 25 dB, the dynamic range of the detected signal can be quite high,e.g., 75 dB. For example, assuming a conventional −20 dBm. If level at1.5 GHz with a 25 dB on-chip programmable attenuator and a requirementthat the side bands be suppressed by 50 dB, the signal strengthindicator 106 should be sensitive to signals from −20 dBm to −95 dBm. Inorder to measure such a varying signal with consistent accuracy thesignal strength indicator 106 is preferably implemented as a logamplifier/detector, as explained in greater detail below.

Third, a problem may exist since various undesired frequency componentsare present. These are all detected at the same time but controlled bydifferent mechanisms. This is preferably solved by using a sharp bandpass filter as described previously, followed by a limiter for the logdetector. The band pass filter 104 pre-selects the frequency componentfor minimization and the limiter effectively eliminates the otherundesired signals that can reduce the measurement accuracy. The limitercaptures only the slightly stronger signals when two or more signals arepresent.

FIG. 8 is a circuit diagram of the preferred embodiment of the envelopedetector 102. The circuit, which is based in part on a Gilbert cellprovides considerably more dynamic range and superior signal to noiseratio than conventional envelope detectors, such as diode-baseddetectors. The circuit includes an upper tree 114 which includes twodifferential transistor pairs 116 and 118, comprising transistorsQ_(2A), Q_(3A), Q_(2B) and Q_(3B). The circuit 110 also includes a lowertree 120 which includes a second set of transistors Q_(1A) and Q_(1B).The upper tree 114 is connected to the lower tree 120, as shown.

A differential input signal is applied directly to the bases of thetransistors of the upper tree 114 at input terminals V_(in) ⁺ and V_(in)⁻. The differential input signal is highly attenuated by a resistordivider network R_(2A),R_(3A), and R_(2B),R_(3B) and fed to the bases ofthe transistors of the lower tree 120. The lower tree 120 is highlydegenerated in comparison to the upper tree 114. The upper treetransistors switch hard while the degenerated lower tree transistors seethe input signal as well as its envelope. Referring back to FIG. 3 theupper tree only sees the zero crossings of the signal 40 and not awareof the envelope 42. The lower tree sees the whole signal 40.

Therefore, the upper and lower trees 114, 120 are used as a multiplier.In this mode, the input signal applied to the upper tree 114 has asignal level exceeding the threshold voltage V_(T) of the transistors(typically about 4 V_(T), where V_(T)≈25 mV_(pp)) whereas the amplitudeof the input signal applied to the degenerate lower tree 120 isconsiderably lower than the threshold voltage (assuming that R₄ is zeroohms) due to R_(2A), R_(3A) and R_(2B), R_(3B) attenuators. The uppertree transistors are saturated and thus switch hard such that currentflows through one side of the upper tree or the other, depending on thepolarity of the input signal. This is schematically represented bysquare wave train 122. In contrast, the transistors of the degeneratedlower tree 120 do not switch hard, and the lower tree functions as anamplifier which is further linearized by the presence of R₄, whereby thecurrents in the collects of Q_(1A) and Q_(1B) are reproductions of thevoltages applied to the bases thereof. [This is schematicallyrepresented by sinusoidal signal 124. However, as a result of the hardswitching of the upper tree transistors the currents in the collectorsof Q_(1A) and Q_(1B) are chopped off. At the output, the collectors ofQ_(2A), Q_(3A) and Q_(2B), Q_(3B) all have a positive polarity, thuseffectively multiplying the input signed by a synchronors square wave,as schematically represented by signal 126.

A set of low pass filters, comprisingR_(6A)C_(2A);R_(5A)C_(1A)+C_(1B);R_(5B),C_(1A)+C_(1B); and R_(6B),C_(2B)average the result so that at V_(out) ⁺ and V_(out) ⁻ the output is adifferential low frequency signal 42 as shown in FIG. 3. Consequently,the output of the circuit 102 represents the envelope of the inputsignal. The net result is that the circuit shifts the spectral contentof the input signal such that the frequency of the desired component,the LSB term in this example, is down converted to zero IF. Thecombination of R_(1A), R_(2A), R_(3A) and R_(1B), R_(2B), R_(3B) plusR₄, not part of a conventional Gilbert cell, provide simultaneousbiasing for the upper tree and biasing plus attenuation for the lowertree for the optimal synchronous detector operation.

FIG. 9 is a circuit diagram of the preferred embodiment of the signalstrength indicator 106. The circuit employ a series of cascadingamplifiers 130, 132 as known in the art per se to provide an outputsignal which is substantially equivalent to the log of the input signal.This enables a linear range of values, e.g. 1 to 5 volts, to representvariations in the level of the input signal on the order of 10⁵. A lowpass filter 134 is incorporated in the feedback path in conjunction withan additional amplifier 136. The circuit effectively provides the bandpass filter 104 of FIG. 5 with the low corner frequency beingestablished by the values of R and C, and the high corner frequencybeing set by the bandwidth of the cascaded series of amplifiers. Thisresults in a first order band pass filter. Higher order tunable activefilters could also be placed between the synch detector of FIG. 9 andthe secondary detector.

The preferred embodiment has been shown and described as operating withdifferential signals. Those skilled in the art will understand that thepreferred embodiment may be readily varied to operate with single endedsignals. Similarly, numerous other modifications may be made to theembodiments described herein without departing from the spirit of theinvention.

1. A quadrature modulator, comprising: a) an in-phase modulation branchreceiving as an input an analog in-phase base band signal, the in-phasemodulation branch including a first dc offset adjustment circuit, afirst base band gain adjustment circuit, and a first mixer; b) aquadrature modulation branch receiving as an input an analog quadraturebased band signal, the quadrature modulation branch including a seconddc offset adjustment circuit, a second base band gain adjustmentcircuit, and a second mixer; c) a local oscillator means for providing alocal oscillator signal to the first mixer and a phase shifted versionof the local oscillator signal to the second mixer; d) a summer forsumming the outputs of the first and second mixers; e) an envelopedetector for detecting an output signal of the modulator and providing asignal representative of the amplitude of the output signal of thequadrature modulator; f) a band pass filter for filtering the amplitudesignal; and g) a signal strength indicator circuit for measuring thestrength of the filtered amplitude signal, the indicator circuitproviding a compensation signal for adjusting the phase shift of thelocal oscillator and the dc offsets and base band gains of the in-phaseand quadrature base band signals.
 2. The modulator according to claim 1,wherein the envelope detector comprises a synchronous detector and thesignal strength indicator comprises a log indicator.
 3. The modulatoraccording to claim 2, including a programmable attenuator for adjustingthe level of the output signal of the quadrature modulator, and whereinthe envelope detector measures the output signal following attenuation.4. The modulator according to claim 1, including a tone generator forsupplying a test tone signal to the in-phase modulation branch input anda ninety degree phase-shifted version of the test tone signal to thequadrature modulation branch input.
 5. The modulator according to claim4, further comprising means for: a) applying a first test tone signal tothe in-phase modulation branch input and a ninety degree phase-shiftedversion of the first test tone signal to the quadrature modulationbranch input; b) employing the compensation signal to minimize carrierleakage in the output signal by adjusting the base band dc offsets inthe in-phase and quadrature branches; c) applying a second test tonesignal to the in-phase modulation branch input and a ninety degreephase-shifted version of the second test tone signal to the quadraturemodulation branch input, wherein the second test tone has a frequencythat is substantially one half of the frequency of the first test tone;and d) employing the compensation signal to minimize an undesired uppersideband frequency component in the output signal by adjusting the baseband gains the in-phase and quadrature modulation branches and the phaseshift of the local oscillator signal.
 6. A method of calibrating aquadrature modulator, comprising: a) applying a first test tone signalto an in-phase modulation branch input of the modulator and a ninetydegree phase-shifted version of the first test tone signal to aquadrature modulation branch input of the modulator; b) measuring thelevel of a local oscillator (LO) feed through in an output signal of themodulator and in response adjusting base band dc offset voltages tominimize the LO feed through; c) applying a second test tone signal tothe in-phase modulation branch input of the Quadrature modulator and aninety degree phase-shifted version of the second test tone signal tothe quadrature modulation branch input; and d) measuring the level of anundesired upper sideband frequency component in the output signal and inresponse adjusting base band gains the in-phase and quadraturemodulation branches and a LO phase error to minimize the undesiredsideband.
 7. The method according to claim 6, wherein the second testtone has a frequency that is substantially one half of the frequency ofthe first test tone.
 8. The method according to claim 6, whereinmeasuring the level of the local oscillator (LO) feed through or the USBin the output signal is carried out by: a) shifting the frequencyspectrum of the output signal such that a lower sideband frequencycomponent (LSB) is down-converted to zero IF; b) filtering thespectrum-shifted signal to pass through either the LO feed through orthe USB; and c) measuring the amplitude of the filtered,spectrum-shifted signal.
 9. The method according to claim 8, wherein thefrequency spectrum of the output signal is shifted by a synchronousenvelope detector.
 10. The method according to claim 9, wherein thesynchronous envelope detector comprises: a) a Gilbert cell having atleast one differential transistor pair in an upper branch and at leastone transistor in a lower branch, the upper and lower branches beinginterconnected, each of the upper and lower branches having inputterminals; b) a resistor divider network connected between the inputterminals of the upper branch and the input terminals of the lowerbranch, the resistive values of the network being selected such that aselected input signal having a signal level sufficient to saturate thetransistors of the upper branch is attenuated so as to not saturate thetransistors of the lower branch; and c) low pass filter means connectedto the upper branch of transistors, an output signal of the detectorbeing provided at the low pass filter.
 11. The method according to claim8, wherein the amplitude of the filtered, spectrum-shifted signal ismeasured by a log detector which provides a compensation signal employedto minimize the LO feed through or undesired sideband.
 12. The methodaccording to claim 11, further comprising selectively attenuating theoutput signal prior to the step of measuring the output signal.
 13. Themethod according to claim 11, wherein the second test tone has afrequency that is substantially one half of the frequency of the firsttest tone.
 14. A quadrature modulator, comprising: a) an in-phasemodulation branch receiving as an input an analog in-phase base bandsignal, the in-phase modulation branch including a first dc offsetadjustment circuit, a first base band gain adjustment circuit, and afirst mixer; b) a quadrature modulation branch receiving as an input ananalog quadrature based band signal, the quadrature modulation branchincluding a second dc offset adjustment circuit, a second base band gainadjustment circuit, and a second mixer; c) a local oscillator means forproviding a local oscillator signal to the first mixer and a phaseshifted version of the local oscillator signal to the second mixer, d) asummer for summing the outputs of the first and second mixers; e)envelope detection means for detecting an output signal of the modulatorand providing a signal representative of the amplitude of the outputsignal of the quadrature modulator; f) band pass filter means forfiltering the amplitude signal; and g) a log detector for measuring thestrength of the filtered amplitude signal, the log detector providing acompensation signal for adjusting the phase shift of the localoscillator and the dc offsets and base band gains of the in-phase andquadrature base band signals.
 15. The modulator according to claim 14,further comprising calibration means for: a) applying a first test tonesignal to the in-phase modulation branch input and a ninety degreephase-shifted version of the first test tone signal to the quadraturemodulation branch input; b) employing the compensation signal tominimize carrier leakage in the output signal by adjusting the base banddc offsets in the in-phase and quadrature branches; c) applying a secondtest tone signal to the in-phase modulation branch input and a ninetydegree phase-shifted version of the second test tone signal to thequadrature modulation branch input, wherein the second test tone has afrequency that is substantially one half of the frequency of the firsttest tone; and d) employing the compensation signal to minimize anundesired upper sideband frequency component in the output signal byadjusting the base band gains the in-phase and quadrature modulationbranches and the phase shift of the local oscillator signal.